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Miller effect

The Miller effect is an electronic phenomenon observed in inverting voltage amplifiers, where the effective capacitance between the input and output terminals appears significantly larger at the input due to the amplifier's voltage gain, thereby influencing the circuit's input impedance and frequency response.^[1] This effect, first described by American electrical engineer John M. Miller in his 1920 paper on the input impedance of three-electrode vacuum tubes, arises from the feedback path formed by the bridging impedance, typically a parasitic capacitance.^[1] In essence, for an amplifier with voltage gain A_v (where A_v is negative for inverting configurations), a bridging capacitor C manifests as an equivalent input capacitance of C(1 - A_v), which can multiply the capacitance by factors exceeding 100 in high-gain stages, limiting bandwidth and requiring design compensations such as cascode configurations.^[2]^[3] Miller's theorem formalizes this by replacing the bridging impedance Z with equivalent impedances at the input and output ports: the input equivalent is Z / (1 - A_v), and the output equivalent is Z / (1 - 1/A_v), simplifying analysis of feedback circuits.^[2] This theorem applies broadly to resistors, inductors, and capacitors, but its most notable impact occurs with capacitors in transistor or op-amp amplifiers, where it reduces high-frequency performance by increasing the time constant at the input.^[3] Originally identified in vacuum tube circuits, the effect persists in modern solid-state devices like CMOS inverters^[4] and FinFETs,^[5] often necessitating techniques such as internal compensation in operational amplifiers to stabilize gain.^[6] Despite its challenges, the Miller effect can be harnessed beneficially in applications like capacitance multipliers, where small physical capacitors are made to behave as much larger ones for filtering or tuning purposes.^[3]

Fundamentals

Definition

The Miller effect refers to the phenomenon in which the capacitance between the input and output of an inverting voltage amplifier appears substantially increased when observed from the input, due to the amplifier's action amplifying the voltage swing across that capacitance. This effective enlargement of the input capacitance stems from the feedback nature of the connection, where changes in input voltage induce larger opposing changes at the output, thereby charging and discharging the capacitance more extensively. The effect was first described in the context of vacuum tube circuits, where it significantly influences input impedance.^[1] In electronics, the Miller effect is primarily observed in active devices such as vacuum tubes, bipolar junction transistors (BJTs), metal-oxide-semiconductor field-effect transistors (MOSFETs), and operational amplifiers (op-amps), arising from either parasitic capacitances inherent to the device structure or intentional feedback elements. These capacitances, typically small in isolation, become problematic at high frequencies by shunting the input signal path and degrading overall circuit performance. For instance, in transistor-based amplifiers, parasitic inter-terminal capacitances couple the input and output, exacerbating the effect in configurations with high gain.^[7] Precisely, the Miller effect manifests as the feedback capacitance C_f appearing multiplied at the input by the factor (1 - A_v), where A_v is the low-frequency voltage gain of the amplifier (typically negative for inverting configurations, yielding an approximate multiplication by |A_v| + 1). A simple illustrative example occurs in a common-source MOSFET amplifier, where the parasitic gate-drain capacitance C_{gd} becomes effectively larger at the gate terminal, often by orders of magnitude depending on the gain, thus dominating the input capacitance and limiting bandwidth.^[7]

Physical Mechanism

The Miller effect arises primarily from the role of a feedback capacitance connected between the input and output terminals of an inverting voltage amplifier. In such a configuration, a small variation in the input voltage, ΔV_in, induces a much larger inverted variation at the output, ΔV_out = -K ΔV_in, where K is the magnitude of the voltage gain (K > 0). Consequently, the voltage swing across the feedback capacitor C_f becomes ΔV_f = ΔV_in - ΔV_out = ΔV_in (1 + K), which is significantly greater than the input swing alone. This amplified voltage difference requires a proportionally larger amount of charge to be supplied through the input to charge the capacitor, effectively increasing the apparent capacitance observed at the input terminal.^[1] This mechanism leads to higher charge accumulation at the input because the charge Q on the capacitor is Q = C_f ΔV_f = C_f (1 + K) ΔV_in. For the input to drive this charge, it must provide an effective capacitance of C_eff = C_f (1 + K), as if the original feedback capacitance were multiplied by the gain factor. In vacuum tube amplifiers, originally analyzed by John M. Miller, this effect was shown to make the apparent input capacity several times greater than the actual inter-electrode capacitances due to the load-dependent feedback. The inversion is crucial, as it adds constructively to the input voltage change, exacerbating the capacitive loading.^[1] In transistor-based amplifiers, the Miller effect manifests through intrinsic capacitances between the control and output terminals, such as the gate-drain overlap capacitance C_gd in MOSFETs or the base-collector junction capacitance C_μ in bipolar junction transistors (BJTs). These parasitic capacitances act as the feedback path, where the high voltage gain from drain/collector to gate/base causes the amplified voltage swing across them, thereby magnifying their impact on the input capacitance and limiting high-frequency performance. For instance, in a common-source MOSFET amplifier, the effective input capacitance contributed by C_gd is C_gd (1 + |A_v|), where A_v is the low-frequency gain, directly tying the physical junction structure to the observed effect.^[8]^[9]

History

Discovery

The Miller effect was first identified by John Milton Miller (1882–1962), an American electrical engineer and physicist who made significant contributions to early radio technology during his tenure at the U.S. National Bureau of Standards.^[10] Born in Hanover, Pennsylvania, Miller earned a Ph.D. in physics from Yale University in 1915 and joined the Bureau as a physicist in 1907, where he conducted pioneering research on vacuum tubes and radio measurements.^[10] In 1919, Miller published his seminal paper titled "Dependence of the Input Impedance of a Three-Electrode Vacuum Tube upon the Load in the Plate Circuit" in the Scientific Papers of the Bureau of Standards, Volume 15.^[11] This work systematically examined how the input impedance of triode vacuum tubes varied with different loads connected to the plate circuit, building on the growing interest in three-electrode tubes for amplification in radio systems following their invention around 1906.^[11] Miller's experiments revealed an unanticipated multiplication of the effective input capacitance under resistive loads, where the grid-to-plate capacitance appeared amplified at the input terminals due to the tube's voltage gain.^[11] Conducted amid efforts to improve radio frequency amplifiers, these observations highlighted how the effect could degrade high-frequency performance by increasing the input susceptance, thereby reducing gain and destabilizing oscillators—critical issues for early wireless communication.^[11] For instance, in tests with a VT-1 tube, the effective grid capacitance rose from 27.9 μμF to 87.6 μμF as the plate load resistance increased from 8,000 ohms to 139,000 ohms, demonstrating the effect's scale in practical setups.^[11]

Early Applications

The Miller effect significantly influenced the design of triode vacuum tube amplifiers during the early 20th century, particularly in radio receivers and transmitters where high-frequency operation was essential. In these circuits, the effect amplified the grid-to-plate capacitance, leading to an increased effective input capacitance that reduced high-frequency gain and limited the overall bandwidth of the amplifier stages. For instance, in grounded-cathode triode configurations common in radio frequency (RF) applications, this feedback mechanism caused energy interchange between the grid and plate circuits, resulting in diminished amplification at frequencies above several kilohertz and complicating signal fidelity in broadcast systems.^[12]^[13] By the 1920s and 1930s, engineers recognized the Miller effect as a key factor in predicting bandwidth limitations for RF amplifiers, integrating it into design guidelines to optimize performance in radio technology. This understanding stemmed from John M. Miller's foundational analysis of input impedance in three-electrode tubes, which highlighted how plate circuit loads altered grid circuit behavior, prompting adjustments in circuit parameters to mitigate frequency distortion. In practice, the effect was quantified in texts like E.L. Chaffee's 1933 treatise, where it was shown to impose constraints on amplification per stage, often limiting gains to less than 2 in RF setups due to interelectrode capacitances of 5-10 pF.^[12]^[13] Specific examples of the Miller effect's impact appeared in tuned circuits and oscillator designs, where it contributed to instability and unwanted regeneration, potentially causing oscillations that disrupted radio signal processing. In tuned RF amplifiers, the amplified capacitance narrowed the bandwidth and altered selectivity, while in oscillators like regenerative circuits, it necessitated careful tuning to avoid self-oscillation. As a direct response, early neutralization techniques emerged in the 1920s, such as Louis Hazeltine's neutrodyne method using balancing condensers to cancel grid-plate coupling, enabling stable multi-stage amplification in receivers without excessive feedback. These approaches, patented by Hazeltine in 1923, became standard in commercial radios by the late 1920s.^[12] The Miller effect continued to pose challenges beyond the vacuum tube era, persisting in early transistor amplifiers of the 1950s where similar feedback capacitances limited high-frequency performance in nascent solid-state radio designs.

Derivation

Basic Amplifier Model

The basic amplifier model for analyzing the Miller effect is an inverting configuration represented by an ideal voltage-controlled voltage source (VCVS) with a voltage gain A_v < 0. This model abstracts the amplifier's behavior, where the input voltage V_{in} applied at the input node produces an output voltage V_{out} = A_v V_{in} at the output node, with both nodes referenced to a common ground. A feedback impedance Z_f connects the input and output nodes, while an input resistor R_{in} is placed in series with the source to drive the input. This setup originated from John M. Miller's investigation of vacuum tube triodes in 1920, where the grid served as the input, the plate as the output, and the filament as ground, with interelectrode capacitance acting as Z_f.^[1] The model assumes an ideal VCVS with infinite input and output impedances, negligible secondary effects, and no additional parasitic components beyond Z_f. Analysis is limited to small-signal AC conditions to linearize the amplifier's response around its operating point. The framework presupposes familiarity with core principles of amplifier voltage gain and network impedance analysis.

Capacitance Calculation

To derive the effective input capacitance due to the Miller effect, consider a basic inverting voltage amplifier model with voltage gain A_v, input voltage V_{in}, output voltage V_{out} = A_v V_{in}, a feedback capacitor C_f connected between the input and output nodes, and a parasitic input capacitance C_{in,parasitic} in parallel with the input. Under the low-frequency approximation, where the gain A_v is constant and frequency-independent, apply Kirchhoff's current law (KCL) at the input node to determine the total input current I_{in}. The current through the parasitic capacitance is I_{parasitic} = j \omega C_{in,parasitic} V_{in}, where \omega is the angular frequency. The current through the feedback capacitor is I_f = j \omega C_f (V_{in} - V_{out}) = j \omega C_f (V_{in} - A_v V_{in}) = j \omega C_f V_{in} (1 - A_v). By KCL, the total input current is the sum: I_{in} = I_{parasitic} + I_f = j \omega C_{in,parasitic} V_{in} + j \omega C_f V_{in} (1 - A_v) = j \omega V_{in} [C_{in,parasitic} + C_f (1 - A_v)]. The effective input admittance is thus Y_{in} = I_{in}/V_{in} = j \omega [C_{in,parasitic} + C_f (1 - A_v)], implying an effective input capacitance of C_{in} = C_{in,parasitic} + C_m, where the Miller capacitance C_m = C_f (1 - A_v). For an inverting amplifier, the voltage gain is negative, A_v = -K with K > 0, so C_m = C_f (1 + K). This shows that the feedback capacitance appears amplified by the factor (1 + K) at the input, significantly increasing C_{in} for high gain K. The derivation assumes negligible loading effects and focuses on the capacitive susceptance, valid at low frequencies where resistive components do not dominate the phase. This formulation was originally validated through measurements on three-electrode vacuum tube circuits, where the computed input capacitance closely matched observed values.^[1]

Effects in Circuits

On Input Impedance

The Miller effect modifies the input impedance of amplifiers by transforming the feedback impedance into an equivalent input impedance that incorporates the voltage gain, particularly impacting high-frequency performance. In John M. Miller's seminal analysis of three-electrode vacuum tubes, the input impedance Z_{in} depends on the load admittance, with the feedback element Z appearing at the input as Z_{in}' = Z / (1 - A_v), where A_v is the internal voltage gain from input to output.^[1] For a feedback capacitance C, this yields an effective Miller capacitance C_m = C (1 - A_v); in inverting configurations where A_v is negative, C_m \approx C (1 + |A_v|), amplifying the capacitance seen at the input by the gain magnitude plus one.^[14] This effective capacitance C_m appears in parallel with the amplifier's intrinsic input resistance R_{in} and any base or gate capacitances, resulting in a total input impedance approximated by

Z_{in} \approx \frac{1}{j \omega C_m + \frac{1}{R_{in}}},

which simplifies to the parallel combination R_{in} \parallel \frac{1}{j \omega C_m}.^[15] At low frequencies, Z_{in} is dominated by R_{in}, but as frequency increases, the j \omega C_m term prevails, causing |Z_{in}| to decrease approximately as $1/(\omega C_m). In real devices, C_m adds to intrinsic capacitances like C_\pi in BJTs or C_{gs} in MOSFETs, yielding a lower high-frequency Z_{in} than expected from low-frequency measurements alone, as the combined capacitance accelerates the impedance roll-off.^[14] In the common-emitter BJT amplifier, the base-collector junction capacitance C_\mu is particularly susceptible to the Miller effect due to the stage's inverting gain A_v = -g_m R_L, where g_m is the transconductance and R_L is the load resistance. This results in C_m \approx C_\mu (1 + g_m R_L), which parallels the base-emitter capacitance C_\pi and input resistance r_\pi, substantially reducing the input impedance at elevated frequencies.^[15] Frequency response plots of |Z_{in}| versus \omega for such amplifiers typically show a flat resistive region at low frequencies, followed by a -20 dB/decade roll-off commencing earlier than in configurations without feedback capacitance; for example, with g_m R_L \approx 50, C_m can exceed C_\pi by an order of magnitude, shifting the corner frequency downward and compressing the range of usable input impedance.^[15] This effect, first detailed in vacuum tube contexts, underscores the need to account for C_m in transistor designs to predict accurate high-frequency behavior.^[1]

On Bandwidth and Gain

The Miller effect introduces a dominant pole in the frequency response of amplifiers, given by f_p = \frac{1}{2\pi R_{in} C_m}, where R_{in} is the input resistance and C_m is the Miller-equivalent capacitance, leading to a gain roll-off of -20 dB per decade beyond f_p.^[16] This pole arises from the amplified feedback capacitance, which increases the effective time constant at the input and dominates the high-frequency behavior in inverting gain stages.^[17] The effective 3-dB bandwidth narrows by a factor of (1 + |A_v|), where A_v is the low-frequency voltage gain, thereby limiting the amplifier's high-frequency operation as gain increases.^[17] For instance, in a common-source or common-emitter amplifier, the 3-dB bandwidth \omega_H is approximated as \omega_H = \frac{1}{R_{out}' (C_{gs} + C_{gd} (1 + |A_{v,LF}|))}, showing the proportional reduction with higher |A_v|.^[17]^[18] In a representative single-stage amplifier simulation, increasing the low-frequency gain |A_v| from 10 to 100 reduces the 3-dB bandwidth by approximately a factor of 10, confirming the inverse proportionality dictated by the Miller-multiplied capacitance.^[17] Such behavior underscores the effect's role in constraining dynamic range for high-gain applications.^[16]

Mitigation Strategies

Neutralization

Neutralization is a classic technique employed to counteract the Miller effect in amplifiers by introducing an auxiliary feedback path that delivers a compensating signal of opposite phase to the voltage across the feedback capacitance C_f. This method, originally developed for vacuum tube circuits, feeds a portion of the output signal back to the input through a dedicated path designed to nullify the degenerative feedback caused by interelectrode capacitances, thereby restoring stability and extending operational bandwidth.^[19] In triode or common-emitter amplifiers, neutralization is implemented by adding a neutralizing capacitor C_n connected from the output (plate or collector) to the input (grid or base), often via an auxiliary coil or transformer winding for phase opposition. The value of C_n is selected such that the effective Miller capacitance becomes C_m = C_f (1 + K) - C_n K \approx 0, where K is the voltage gain of the stage, effectively canceling the amplified input capacitance and mitigating bandwidth reduction. This adjustment ensures that the net feedback current through the input capacitance is minimized, allowing higher gain without oscillation.^[19]^[20] The technique was pioneered by Louis Alan Hazeltine in 1922 and became widely adopted in 1930s radio-frequency (RF) amplifiers, particularly in Neutrodyne receivers, to enable multi-stage amplification for broadband operation without parasitic oscillations. By the end of the decade, it facilitated the design of stable tuned RF stages in early broadcast receivers, though it was eventually supplanted by tubes with inherently lower feedback capacitances.^[20]^[21] The neutralization technique continues to be used in modern transistor-based applications, such as RF power amplifiers, to suppress parasitic oscillations.^[22] Despite its effectiveness, neutralization is sensitive to precise tuning of C_n, as mismatches can lead to incomplete cancellation or even instability. It performs best at a single operating frequency, where the phase relationships are fixed, but exhibits residual effects in broadband applications due to frequency-dependent variations in gain and parasitic elements.^[19]

Cascode Configuration

The cascode configuration is a multi-transistor amplifier topology that stacks a lower transistor in common-source (for MOSFETs) or common-emitter (for BJTs) configuration with an upper transistor in common-gate or common-base configuration, respectively. This arrangement isolates the input of the lower transistor from the large output voltage swings at the load, as the source (or emitter) of the upper transistor presents a low-impedance node that buffers the intermediate connection.^[23]^[24]^[25] The mechanism for minimizing the Miller effect lies in the reduced voltage gain across the feedback capacitances. For the lower transistor's gate-drain (or base-collector) capacitance C_{gd1} (or C_{\mu1}), the voltage swing at the drain (or collector) is minimized by the current-buffering action of the upper transistor, resulting in an effective capacitance of approximately $2C_{gd1} rather than the amplified (1 + |A_v|)C_{gd1} seen in a single-stage amplifier. The upper transistor's C_{gd2} contributes negligibly to the input, appearing as C_{gd2}/A_{v1} at the overall input, where A_{v1} is the low-frequency gain of the lower stage. Thus, the total effective Miller capacitance is roughly C_{gd1} + C_{gd2}/A_{v1}.^[23]^[25] This reduction yields key advantages, including bandwidth extension by factors of 10 to 100 times in radio-frequency applications through decreased input capacitance and pole shifting, enhanced reverse isolation between input and output ports (often >60 dB), and improved linearity by curtailing feedback-induced distortion.^[26]^[23] In practical examples, BJT cascodes are utilized in audio amplifiers to preserve signal fidelity across the audible spectrum by suppressing parasitic feedback, while MOSFET cascodes dominate RF low-noise amplifiers (LNAs) for wideband operation up to several GHz.^[25]^[26] For an ideal cascode, the bandwidth improvement stems from the lower stage's gain A_{v1} \approx g_{m1} R_L', where R_L' is the effective load on the lower stage (approximately $1/g_{m2}); this shifts the 3-dB frequency f_{3\text{dB}} from roughly $1/(2\pi R_L C_{gd1}) in a single stage to near the transistor's transition frequency f_T \approx g_{m1}/(2\pi C_{gs1}), yielding a multiplicative factor approaching A_{v1}.^[23]

Miller's Theorem

General Statement

Miller's theorem generalizes the Miller effect observed in vacuum tube amplifiers to arbitrary impedances connected between the input and output ports of a linear, unilateral voltage amplifier. Originally derived in the context of triode tube input capacitance, the theorem allows the bridging impedance to be replaced by two equivalent impedances: one in shunt with the input and one in shunt with the output, without altering the overall voltage and current behavior at those ports. This simplification facilitates circuit analysis, particularly for feedback networks in amplifiers.^[13]^[27] Formally, consider an impedance Z connected between the input node at voltage V_1 and the output node at voltage V_2 = A V_1, where A is the complex voltage gain of the amplifier from input to output. The theorem states that this impedance is equivalent to an input impedance Z_1 = \frac{Z}{1 - A} appearing across the input terminals and an output impedance Z_2 = \frac{Z}{1 - 1/A} appearing across the output terminals. These equivalents maintain the same terminal voltages and currents as the original configuration.^[27]^[28] The proof proceeds from basic circuit principles of voltage division and current conservation, assuming the amplifier transmits signals only in the forward direction (unilaterality). The voltage across Z is V_1 - V_2 = V_1 (1 - A), so the current through Z is I_Z = \frac{V_1 (1 - A)}{Z}. At the input, this current appears as if supplied by an equivalent impedance Z_1, satisfying I_Z = \frac{V_1}{Z_1}, which directly yields Z_1 = \frac{Z}{1 - A}. For the output equivalent, the current I_Z is viewed from the output perspective: I_Z = \frac{V_2 - V_1}{Z} = \frac{A V_1 - V_1}{Z} = \frac{V_1 (A - 1)}{Z}, or equivalently I_Z = \frac{V_2}{Z_2'} where Z_2' is the series combination leading to the shunt Z_2 = \frac{Z}{1 - 1/A}, ensuring current balance when the input is driven. This derivation holds for any linear impedance Z, including resistors, capacitors, or inductors, provided $1 - A \neq 0 and A \neq 0.^[27]^[29] A key special case recovers the original Miller effect when Z = \frac{1}{j \omega C} is a capacitor, resulting in an equivalent input capacitance C_1 = C (1 - A). For inverting amplifiers, where A is negative (e.g., A = -|A| with |A| \gg 1), this simplifies to C_1 \approx C |A|, demonstrating the apparent multiplication of capacitance at the input due to feedback. The theorem's approximations are most accurate when |A| \gg 1, and it assumes no loading effects from the replacement impedances themselves.^[13]^[27]

Applications and Approximations

In high-gain amplifiers, such as operational amplifiers (op-amps) and multi-stage configurations, Miller's theorem provides a key approximation when the voltage gain |A| is large. The input impedance Z_{\text{in}} seen across the bridging impedance Z simplifies to Z_{\text{in}} \approx -A Z, where the negative sign arises from the inverting nature of typical amplifiers. This approximation facilitates hand analysis by treating the effective input impedance as dominated by the amplified feedback element, effectively reducing the perceived input resistance to a fraction of Z (e.g., Z_{\text{in}} \approx Z / (1 + K) for gain -K, approaching near-zero for infinite K). For capacitive Z = 1/(Cs), this manifests as an effective input capacitance C_{\text{in}} \approx C(1 + |A|), which is crucial for estimating frequency response without full circuit simulation.^[3] This approximation finds extensive use in circuit design for predicting stability in feedback loops and analyzing parasitic effects in integrated circuits (ICs). In multi-stage op-amps, Miller's theorem models compensation capacitors to split poles, creating a dominant low-frequency pole that enhances phase margin (e.g., up to 90° in three-stage designs with indirect compensation). It aids in forecasting right-half-plane zeros from feed-forward parasitics, which degrade stability, allowing designers to adjust capacitor sizes (e.g., reducing from 10 pF to 1-2 pF) for robustness against process variations up to 50%. In ICs, the theorem quantifies parasitic capacitances (e.g., gate-drain C_{\text{gd}}) in feedback paths, enabling accurate pole-zero placement via two-port analysis to avoid inaccuracies in traditional DC-gain assumptions.^[30]^[31] Modern extensions of Miller's theorem apply to power electronics, particularly in widebandgap devices like GaN and SiC switches, where the Miller plateau—a flat gate-source voltage region during switching—limits dV/dt and risks unintended turn-on. In enhancement-mode GaN FETs, the theorem highlights how gate-drain capacitance amplifies transients, with peak dV_{ds}/dt reaching values up to those tested at 400 V (e.g., influenced by gate resistance from 0 to 20 Ω), necessitating refined models beyond conventional plateau equations for fast switching. For SiC MOSFETs in half-bridge topologies, Miller current injection via C_{\text{GD}} causes V_{\text{GS}} spikes exceeding thresholds (e.g., 3.2 V), but active Miller clamps shunt this current, reducing glitches (e.g., from +2.4 V to -3.5 V) and enabling dV/dt up to 90 V/ns safely. Post-2020 trends emphasize resonant gate drivers for GaN/SiC, recovering gate charge to cut losses by 26% at 2.5 MHz, aligning with efficiency demands in high-voltage applications.^[32]^[33]^[34] In RF low-noise amplifiers (LNAs), Miller's theorem supports broadband matching by mitigating gate-drain capacitance effects in cascode topologies, enhancing noise figure (NF) and gain flatness across wide bands. For instance, graded-channel GaN HEMT cascodes in Ka-band (20-40 GHz) reduce the Miller effect compared to common-source stages, achieving NF below 1.5 dB (down to ~1 dB at 30-37 GHz) with 15 ± 1 dB gain via high-pass RC networks and feedback. Reactive Miller-effect filters resonate input impedances over the band, minimizing gain ripples and intermodulation while ensuring unconditional stability. To illustrate, consider estimating effective capacitance: in a single-stage common-emitter amplifier, a gate-drain capacitance C_{\mu} yields C_{\text{eff}} = C_{\mu} (1 + g_m R_C), potentially limiting bandwidth; in a cascode (common-emitter + common-base), the Miller multiplication is minimized to near C_{\mu}, preserving high-frequency performance without direct input-output coupling.^[35]^[36]^[37]

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This effect, now commonly known as the Miller Effect, was first reported by John Miller in the following paper. Consider the following amplifier with voltage ...
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The latter capacitance (between base and collector) gives rise to so called Miller effect. Due to Miller effect the amplifiers with increased gain demonstrate.<|control11|><|separator|>
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[33]
Ultra‐Low Noise Figure Ka‐Band MMIC LNA With Graded‐Channel ...
Apr 27, 2025 · The cascode LNA also has a reduced Miller effect which helps with overall noise performance and has higher gain [5, 6] than a common source ...Missing: post- | Show results with:post-
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Cascode Amplifier Configuration: Advantages and Disadvantages
Oct 11, 2023 · A cascode configuration reduces the Miller capacitance effect and improves stability by minimizing effective input capacitance. Improved ...